Controlled Magnetic Resonance in High Efficiency High Frequency Resonant Power Conversion

ABSTRACT

A method includes magnetically coupling a first coil of a wireless power transfer system to a second coil of the wireless power transfer system, wherein the first coil is coupled to a first resonant tank comprising a first variable capacitance network, and the second coil is coupled a second resonant tank comprising a second variable capacitance network, and adjusting a capacitance of at least one of the first variable capacitance network and the second variable capacitance network to regulate an output voltage of the wireless power transfer system.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of and claims the benefit of U.S.patent application Ser. No. 15/657,500, filed on Jul. 24, 2017, andentitled “High Efficiency High Frequency Resonant Power Conversion,”which is a continuation of and claims the benefit of U.S. patentapplication Ser. No. 14/177,049, filed on Feb. 10, 2014, and entitled“High Efficiency High Frequency Resonant Power Conversion,” now U.S.Pat. No. 9,755,534 issued Feb. 10, 2014, which claims priority to U.S.Provisional Application No. 61/850,423, Feb. 14, 2013, and entitled“High Efficiency High Frequency Resonant Power Conversion,” whichapplications are incorporated herein by reference.

TECHNICAL FIELD

The present invention relates to power converters, and, in particularembodiments, to resonant power conversion suitable for high efficiencyand high frequency operations.

BACKGROUND

A power supply is used to convert electric power from one form toanother to suit the particular load in the system. Generally, there is astrong desire to operate the power supply at a high switching frequencyto reduce the size and improve the performance of the power supply. Ahigh frequency operation is especially important for applications inwireless power transfer, where higher frequency helps also to transfermore power over longer distance. For example, many wireless powertransfer systems operate at 6.78 MHZ or 13.57 MHZ.

To reduce the power loss of power switches at high switching frequency,resonant power converters were developed to improve the efficiency ofthe power conversion. However, a resonant power converter can achievethe best efficiency only when it operates at or close to a resonantfrequency. Because of the component tolerance of the resonant tank, theresonant frequency cannot be easily set to a particular value, so it isdifficult to maintain high efficiency operation of a resonant converter.

Also, in many wireless power transfer systems, there is a need tocontrol the voltage and power at the receiver. Currently a separatepower stage is used to regulate the power transfer. Such a system incurshigh cost and suffers from low efficiency.

SUMMARY OF THE INVENTION

These and other problems are generally solved or circumvented, andtechnical advantages are generally achieved, by preferred embodiments ofthe present invention which provides an improved resonant powerconversion.

In accordance with an embodiment, an apparatus comprises a first switchnetwork comprising a plurality of power devices coupled to a first porthaving a first voltage, and a first resonant tank coupled to the firstswitch network, wherein the first resonant tank includes a first coiland a first variable capacitance network comprising a first capacitor inseries with a first variable capacitor network, and wherein the firstvariable capacitor network comprises a voltage shifting diode inparallel with a plurality of capacitor-switch branches, and wherein eachcapacitor-switch branch comprises a capacitor connected in series with acontrol switch.

In accordance with another embodiment, a method comprises magneticallycoupling a first coil of a wireless power transfer system to a secondcoil of the wireless power transfer system, wherein the first coil iscoupled to a first resonant tank comprising a first variable capacitancenetwork, and the second coil is coupled a second resonant tankcomprising a second variable capacitance network, and adjusting acapacitance of at least one of the first variable capacitance networkand the second variable capacitance network to regulate an outputvoltage of the wireless power transfer system.

In accordance with yet another embodiment, a system comprises a firstcoil of a power system magnetically coupled to a second coil of a powersystem, wherein the first coil is coupled to a first resonant tankcomprising a first variable capacitance network, and the second coil iscoupled to a second resonant tank comprising a second resonantcapacitor, and a controller configured to regulate an output voltage ofthe power system through adjusting a capacitance of the first variablecapacitance network.

The foregoing has outlined rather broadly the features and technicaladvantages of the present invention in order that the detaileddescription of the invention that follows may be better understood.Additional features and advantages of the invention will be describedhereinafter which form the subject of the claims of the invention. Itshould be appreciated by those skilled in the art that the conceptionand specific embodiment disclosed may be readily utilized as a basis formodifying or designing other structures or processes for carrying outthe same purposes of the present invention. It should also be realizedby those skilled in the art that such equivalent constructions do notdepart from the spirit and scope of the invention as set forth in theappended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention, and theadvantages thereof, reference is now made to the following descriptionstaken in conjunction with the accompanying drawings, in which:

FIG. 1 illustrates a power stage schematic diagram of a resonant powerconverter;

FIG. 2 illustrates a schematic diagram of a first illustrativeembodiment of a resonant power converter in accordance with variousembodiments of the present disclosure;

FIG. 3 illustrates the key waveforms of the resonant power convertershown in FIG. 2 in accordance with various embodiments of the presentdisclosure;

FIG. 4 illustrates a schematic diagram of an embodiment of a resonantpower converter in accordance with various embodiments of the presentdisclosure;

FIG. 5 illustrates a diagram of an embodiment of a resonant convertershown in FIG. 4 with a control system in accordance with variousembodiments of the present disclosure;

FIG. 6 illustrates the key waveforms of the resonant power convertershown in FIG. 5 in accordance with various embodiments of the presentdisclosure;

FIG. 7 illustrates an embodiment of a resonant power converter withvarious embodiments of the present disclosure;

FIG. 8A illustrates an embodiment of a variable capacitance inaccordance with various embodiments of the present disclosure;

FIG. 8B illustrates an embodiment of a control mechanism of a variablecapacitance in accordance with various embodiments of the presentdisclosure;

FIG. 8C illustrates an embodiment of an implementation of abidirectional switch in a variable capacitance in accordance withvarious embodiments of the present disclosure;

FIG. 9A illustrates an embodiment of a capacitor in accordance withvarious embodiments of the present disclosure;

FIG. 9B illustrates an embodiment of a voltage shifting techniqueapplied to the capacitor in FIG. 9A in accordance with variousembodiments of the present disclosure;

FIG. 9C illustrates another embodiment of a voltage shifting techniquein accordance with various embodiments of the present disclosure;

FIG. 10 illustrates embodiments of a resonant power converter with thevoltage shifting technique shown in FIG. 9C in accordance with variousembodiments of the present disclosure;

FIG. 11 illustrates key waveforms of a resonant power converter in FIG.10 in accordance with various embodiments of the present disclosure;

FIG. 12A illustrates an embodiment of a configuration of a variablecapacitance with unidirectional switches in accordance with variousembodiments of the present disclosure;

FIG. 12B illustrates an embodiment of a control mechanism of a variablecapacitance with unidirectional switches in accordance with variousembodiments of the present disclosure;

FIG. 12C illustrates an embodiment of a unidirectional switch of avariable capacitance with unidirectional switches in accordance withvarious embodiments of the present disclosure;

FIG. 13 illustrates an embodiment of a Class-E power converter with thevariable capacitance technique shown in FIG. 12A in accordance withvarious embodiments of the present disclosure;

FIG. 14 illustrates an embodiment of a wireless power transfer systemwith variable capacitance technique in accordance with variousembodiments of the present disclosure;

FIG. 15A illustrates an embodiment of a system block diagram of aresonant power converter with capacitance control in accordance withvarious embodiments of the present disclosure;

FIG. 15B illustrates an embodiment of a relationship between an outputvoltage and a transmitter capacitance in a resonant power converter withcapacitance control in accordance with various embodiments of thepresent disclosure;

FIG. 15C illustrates an embodiment of a relationship between an outputvoltage and a receiver capacitance in a resonant power converter withcapacitance control in accordance with various embodiments of thepresent disclosure;

FIG. 16 illustrates an embodiment of a resonant power converter withcapacitance control for wireless power transfer without usingcommunication between the transmitter and the receiver in voltagecontrol in accordance with various embodiments of the presentdisclosure;

FIG. 17 illustrates an embodiment of a resonant power converter withcapacitance control for wireless power transfer using communicationbetween the transmitter and the receiver in voltage control inaccordance with various embodiments of the present disclosure;

FIG. 18 illustrates an embodiment of a wireless power transfer systemwith one transmitter coupled to multiple receivers in accordance withvarious embodiments of the present disclosure;

FIG. 19 illustrates an embodiment of a current sense technique inaccordance with various embodiments of the present disclosure;

FIG. 20 illustrates an embodiment of another current sense technique inaccordance with various embodiments of the present disclosure;

FIG. 21 illustrates an embodiment of a gate drive technique withnegative voltage ability in accordance with various embodiments of thepresent disclosure;

FIG. 22 illustrates the key waveforms of the gate drive technique shownin FIG. 22 in accordance with embodiments of the present disclosure;

FIG. 23 illustrates an embodiment of a gate drive technique withpositive feedback in accordance with various embodiments of the presentdisclosure;

FIG. 24 illustrates an embodiment of integrated implementation of aresonant power converter in accordance with various embodiments of thepresent disclosure;

FIG. 25 illustrates another embodiment of integrated implementation of aresonant power converter in accordance with various embodiments of thepresent disclosure, and

FIG. 26 illustrates yet another embodiment of integrated implementationof a resonant power converter in accordance with various embodiments ofthe present disclosure.

Corresponding numerals and symbols in the different figures generallyrefer to corresponding parts unless otherwise indicated. The figures aredrawn to clearly illustrate the relevant aspects of the variousembodiments and are not necessarily drawn to scale.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

The making and using of the presently preferred embodiments arediscussed in detail below. It should be appreciated, however, that thepresent invention provides many applicable inventive concepts that canbe embodied in a wide variety of specific contexts. The specificembodiments discussed are merely illustrative of specific ways to makeand use the invention, and do not limit the scope of the invention.

The present invention will be described with respect to preferredembodiments in a specific context, namely in a resonant power converteror resonant power conversion system. The invention may also be applied,however, to a variety of power converters including various isolatedpower converters such as full-bridge converters, half-bridge converters,forward converters, flyback converters and/or the like, non-isolatedpower converters such as buck converters, boost converters, buck-boostconverters and/or the like, any combinations thereof and/or the like.Hereinafter, various embodiments will be explained in detail withreference to the accompanying drawings.

Resonant power converters have been used for a long time to increase theefficiency of power converters at high switching frequencies. There aremany resonant converter topologies. FIG. 1 shows a resonant powerconverter, consisting of a switch network 101 (S1 through S4), aresonant tank 102 (Lr and Cr), power transformer T, and rectifier 103(D1 through D4). The resonant inductor Lr can be the leakage inductanceof transformer T, or a discrete inductor, or any combination thereof. Cris the resonant capacitor. The power transformer T has np turns ofprimary winding, and ns turns of secondary winding. S1 through S4 arethe primary switches, and D1 through D4 form the secondary rectifier. Itis well known that a diode in the secondary rectifier can be replaced bya synchronous rectifier, i.e. an active switch such as a power MOSFETwith a proper drive to operate similarly to a diode. Co is the outputcapacitor, and Ro represents the load, which can be an actual load suchas a power input to integrated circuits, or one or more power converterscoupled to one or more actual loads, such as in a battery chargecircuit. Vin represents the input power source. The resonant capacitorCr and resonant inductor Lr form a resonant tank with a resonantfrequency fr. It is known that fr=1/(2π√{square root over (LrCr)}),where Lr is the inductance of resonant inductor Lr, and Cr is thecapacitance of resonant capacitor Cr.

It is well known that a resonant converter can achieve high efficiencyat or around its resonant frequency. Another interesting feature is thatat the resonant frequency, the voltage gain of a resonant converter isapproximately equal to 1 regardless of the load, if the power losses ofthe components can be ignored as in most high-efficiency powerconverters. In such a case, for the converter with a full-bridge primaryswitch network and full-bridge secondary rectifier as shown in FIG. 1,the output voltage is basically determined by the turns-ratio of thetransformer (ignoring the power loss in the converter):

Vo=Vin*ns/np

This relationship shows that it is especially attractive to operate aresonant converter at its resonant frequency in a bus converter, inwhich the output voltage changes proportionally with the input voltage.However, because the actual inductance value of Lr and the actualcapacitance of Cr may vary depending on differences in theirmanufacturing process and their operating conditions, it is impossibleto know the exact resonant frequency of a resonant converter withoutactually measuring it. In the design of such a resonant bus converter,it would be necessary to tightly control the tolerance of resonantcomponents Lr and Cr, but such practice significantly increases thecomponent cost.

The scheme showing in FIG. 2 can force the converter to operate at afrequency substantially equal to its resonant frequency (for examplewithin a range of +/−10%) in steady state. The transformer T has afeedback winding 204 with nf turns. In one embodiment, the feedbackwinding is in the primary side of the transformer. In anotherembodiment, the feedback winding is part of the primary winding. Thevoltage amplitude generated by the feedback winding 204 substantiallyfollows the output voltage, and thus provides a rough feedback signal ofthe output voltage with low cost (no signal is needed to cross theisolation boundary). Using a signal conditioning circuit 205, a feedbacksignal 207 substantially proportional to the output voltage can begenerated. In accordance with an embodiment of this invention, thesignal conditioning circuit performs rectification and filteringfunction. The feedback signal 207 is fed to a regulation circuit 208. Ina preferred embodiment of this invention, the block 208 composes of anerror amplifier and a frequency control block. A signal proportional tothe input voltage (Vin/M) is coupled to the error amplifier as areference, and the feedback signal 207 is coupled to the error amplifieras a feedback signal. In a preferred embodiment, the output of the erroramplifier adjusts the switching frequency of primary switch network 201.As a result, in the steady-state operation the output voltage is inproportional to the input voltage. With a proper selection of nf and M,the voltage gain of the resonant converter can be regulated tosubstantially 1, i.e. the converter operates at a frequencysubstantially the same as the resonant frequency of the resonant tank202 in steady state. The presence of the feedback can also improve theperformance in transients, such as during start-up, or when the load ischanging. To control the output voltage more accurately, more signalconditioning can be added in circuit 205 or the error amplifierreference 206 to compensate the effect of power losses and possibleglitches associated with the switching action of power switches S1through S4. For example, current information (such as the transformercurrent, input current, or a switch current) can be added to thefeedback signal or the reference signal to reduce the output voltagechange caused by load current change, or to intentionally make thevoltage increase or decrease with the load current.

FIG. 3 shows simulated waveforms of the converter in FIG. 2. S1 throughS4 operate with a fixed duty cycle, with a transition in which bothswitches in a leg are turned off to allow the switches be transitionedinto next states with soft switching. The Leg Voltage refers to thevoltage at the junction of the source of S1 and drain of S2. From thesimulation, it can be seen that zero-voltage switching (ZVS) at turn-onhas been achieved for the primary switches. However, by properlyselecting the mutual inductance of the transformer, near zero-currentswitching turn-off can also be achieved for the primary switches. Thesecondary diodes (or synchronous rectifiers if they are used) can beturned off with zero current. Therefore, very high efficiency can beobtained. The reflected voltage is the voltage signal obtained at thefeedback winding 204. It can be seen that the amplitude of the reflectedvoltage is very close to the output voltage. Therefore, the signal 207is approximately proportional to the output voltage throughrectification and filtering, and thus suitable to be used as a feedbacksignal of the output voltage.

It can be seen from FIG. 3 that the reflected voltage has a ripple muchhigher than the ripple of the output voltage. This is due to effect ofthe leakage inductance of the transformer. To reduce such a deviation,the resonant tank can be divided into two, one at the primary side, andone at the secondary side of the transformer, as is shown in FIG. 4. Theresonant tank 402 at the primary side and 410 at the secondary sideshould have substantially the same (for example within +/−10%) resonantfrequency and power capability, i.e. the parameters should follow thefollowing relationship closely:

Cr2=Cr1*(np/ns)²

Lr2=Lr1*(ns/np)²

Please note that if no additionally discrete resonant inductors are usedfor the resonant inductance, the leakage inductances in the primary sideand the secondary side automatically follow the above equation. Theconverter also works well even if the parameters are slightly off thecorrect value. For example, the parameter may deviate from the idealvalues by 10%.

An automatic resonant frequency tracking circuit similar to the oneshown in FIG. 2 can be used to operate this converter at or near theresonant frequency, as shown in FIG. 5. Because Cr2 compensates theleakage inductance of the transformer on the secondary side, thereflected voltage signal has much lower ripple than that shown in FIG.2, and thus can be used to get better control performance. FIG. 6 showsthe key simulated waveforms.

It can be seen that the converter operates similarly to the one in FIG.2, except that the reflected voltage has much less ripple. Thedifference between the output voltage and the reflected voltage inamplitude is mainly due to the power losses in the secondary sidecomponents, because the reflected voltage of the feedback winding 504mainly represents the voltage induced by the transformer mutual flux inthe transformer windings. Usually, such a deviation is acceptable in abus converter. But if a tighter regulation is required due to somereasons, the deviation can be reduced by compensating the power lossesin the signal conditioning block 505 or the reference 506 of the erroramplifier. For example, current information can be used to represent thevoltage drop caused by the load current, and be used to modify a signalin the conditioning block 505, or the reference 506 at the erroramplifier.

The above techniques can have different embodiments. In someembodiments, resonant tanks 202, 502 and 510 can also be a parallelresonant tank. In other embodiments, resonant tanks 202, 502 and 510 canbe a parallel-series resonant tank. In some embodiment, the transformerT can have center-tap secondary windings. In some embodiments, theswitch network and/or the rectifier 201, 203, 501, 503 can use a halfbridge topology. In some embodiments, the switch network and/or therectifier 201, 203, 501, 503 can use a push-pull topology. As long asthe transformer has relatively good coupling between the primary windingand the secondary winding (for example the coupling coefficient ishigher than 0.9), the reflected voltage from the feedback winding is areasonable representation of the output voltage, and the frequencytracking scheme should work well. The frequency control block in FIG. 2and FIG. 5 can have different implementations. A good way is to presetthe switching frequency at a value higher than the possible maximumresonant frequency, and use the feedback mechanism to adjust down thefrequency to the correct value. Alternatively, the switching frequencycan be preset at a value lower than the possible minimum resonantfrequency, and use the feedback mechanism to adjust up the frequency tothe correct value.

This technique can work well with multi-output power converters.Additional secondary windings can be added to the transformer T, witheach winding having its own resonant capacitor and rectifier circuit. Asthe resonant capacitor in each secondary winding compensates its leakageinductance, the cross regulation of the outputs is improved comparedexisting topologies.

The above technique works well if the switching frequency of a powerconverter is allowed to change over a range. However, some applications,such as certain wireless power transfer (WPT) may require the converterto operate at a fixed frequency (such as 6.78 MHz or 13.56 MHz). Becausethe frequency is very high, resonant technologies are preferred for suchapplications. Using resonant technologies can transfer power wirelesslyacross a considerable distance. Resonant wireless power transfer systemswith a high quality factor (high Q) (resonant coupling technique) allowsmore power to be transferred even when the coupling coefficient betweenthe transmitting coil and the receiving coil is small.

FIG. 7 shows a resonant topology which can be used with the resonantcoupled WPT. A half bridge switch network 701 consisting of S1 and S2converts input dc voltage Vin into pulses at a high frequency. Usually,the switch duty cycle is fixed with a transition in which both switchesare turned off to allow the switches be transitioned into next stateswith soft switching. L1 is the transmitter coil (primary winding of atransformer), and L2 is the receiver coil (secondary winding of atransformer). In a normal power converter L1 and L2 are tightly coupled.In a WPT L1 and L2 may be loosely coupled. L1 and L2 can be consideredas a transformer with coupling coefficient k between primary andsecondary coils. Lx1 is the leakage inductance of the transmitter coil,plus any additional inductance such as from a discrete inductor or anyparasitic inductance from connecting traces. Lx2 is the leakageinductance of the receiving coil L2 plus any additional inductance suchas from a discrete inductor or parasitic inductance of connectingtraces. There are two resonant tanks: 702 in the transmitter in whichCr1 is the resonant capacitor, and 710 in the receiver in which Cr2 isthe resonant capacitor. D1 and D2 form a half-bridge rectifier 703 inthe receiver, and convert the ac current in the receiving coil into a dccurrent. Co is the output capacitor for smoothing the output voltage.The load can be an actual load or dc-dc or dc-ac power convertersconnecting to the load.

In the above embodiment a half bridge topology is used in thetransmitter and receiver. In other embodiments, full bridge, push-pull,or Class E topologies, can also be used. However, the half-bridgearrangement in both transmitter and receiver has one prominentadvantage: the resonant capacitors can be connected to the return leadof the input or output. This arrangement allows easier adjustment of thecapacitance of such capacitors. This is particularly important inwireless power transfer systems for mobile devices, because such systemsmay be required to work at a fixed frequency such as 6.78 MHz.

In such fixed frequency systems, it is necessary to adjust the resonantfrequency of the resonant tanks 702 and 710 to achieve desirableperformance. Because it is quite cumbersome to adjust the inductance ofa coil in real time, especially the leakage inductance of a transformer,it is more practical to adjust the capacitance of a resonant capacitor.It is well known for a long time that a variable capacitor can becontrolled to get different capacitance by applying different voltage,and such variable capacitors can be used as the resonant capacitors.However, the power handling capability of variable capacitors is stilllimited at this time. It may be desirable to use switched capacitors toadjust the capacitance, as shown in FIG. 8A, FIG. 8B, and FIG. 8C. FIG.8A shows the configuration of a variable capacitor 810, in which a fixedcapacitor Cr0 and several capacitors Cx1 through Cx4 are included, andswitches Sx1 through Sx4 are used to switch in or switch out capacitorsCx1 through Cx4. In a real system, the number of switches and capacitorscan be adjusted according to design needs and application requirements.The capacitor switches Sx1 through Sx4 are not required to switch athigh frequencies, therefore their design is mainly concerned withconduction loss. If the capacitance of each branch is different, thenthe size and current capability of the switches can be different too,and should be optimized according to the capacitance it controls. FIG.8B shows a control mechanism 820 for variable capacitance. Thecapacitance requirement is represented by a digital signal, and if theinput signal is an analog signal, then a digitizer such as an A/Dconverter should be used to convert it into digital form. A decoder isused to translate the digital signal into switch control signal for theswitches. Although Cx1 through Cx4 can have any capacitance as desired,it would be more advantageous to have a substantially doublingrelationship between these capacitors:

Cx2=2Cx1,Cx3=2Cx2,Cx4=2Cx3

In this way, it's possible to use a limited number of capacitors to getmany different levels of capacitance with substantially equal step size.In a half bridge configuration such as in FIG. 7, the resonant capacitorvoltage may be unidirectional if the quality factor Q of the resonanttank is low. In such a case, a switch can be implemented as aunidirectional device such as MOSFETs, BJT, or IGBT. However, in manyapplications such as in a WPT system, it is desirable to achieve a highquality factor, so the resonant capacitor usually sees a bidirectionalvoltage. The switch device usually has to be implemented as abidirectional device, such as two unidirectional devices (such asMOSFETs) in back-to-back connection as shown in FIG. 8C. Thebidirectional switches and their corresponding complex drive circuitsresult in high cost and high power losses in the design. Therefore, itis highly desirable to avoid the use of bidirectional switches.

FIGS. 9A, 9B and 9C show a way to shift the voltage of resonantcapacitors. For the resonant capacitor Cr in FIG. 9A, a newconfiguration 920 with two capacitors Cr1 and Cr2 in series and aclamping diode across Cr2 can be used as is shown in FIG. 9B. To havethe same equivalent capacitance, the relationship of Cr=Cr1Cr2/(Cr1+Cr2)should be kept. The clamping diode Dc limits the negative voltage of Cr1to zero (actually a diode voltage drop, to be exact), thus capacitor Cr1can be treated as a unidirectional device. Very interestingly, Dcconducts very little current in steady state, and has almost no powerloss and no impact on the resonant process. However, it does shift thedc voltages of Cr1 and Cr2 automatically to lift the lowest point ofCr1's voltage waveform to near zero. To speed up the voltage shifting intransients and set the right dc voltages, a dc current path should beprovided. In one embodiment, a circuit 931 capable of conducting a dccurrent is connected in parallel with a Cr2, and a circuit 932 capableof conducting a dc current is connected in parallel with Cr1. In oneembodiment, circuits 931 and 932 are resistors, as is shown as Rx1 andRx2 in FIG. 9C. The values of Rx1 and Rx2 should be chosen such thatthey do not cause significant power losses. Also, because any capacitorhas an internal parasitic parallel resistance, it is also possible touse the internal resistance to replace the parallel resistor. As aresult, discrete resistors across the resonant capacitors may not benecessary.

FIG. 10 shows the resonant converter in FIG. 7 with the voltage shiftingtechnique. Switching network 1001 is two switches configured in halfbridge, and resonant capacitor networks 1002 and 1010. The output iscoupled to the resonant network through a rectifier 1003. FIG. 11 showsthe key simulated waveforms. It can be seen clearly that the clampingdiode only conducts a small current when the corresponding capacitorvoltage is around its lowest point. This small current can establish theright dc voltages in the capacitors without considerable effect on theresonant process. The amount of current the clamping diode conducts isalso related to the resistances across the capacitors, and the higherthe resistance, the lower the current. It should be noted that, assumingthe ac parts of the voltages across the resonant capacitors aresinusoidal, the dc voltages in the resonant capacitors in thetransmitter in steady state are:

${{Vdc}\; 1\; b} = {{\sqrt{2}Vr1b} = {\sqrt{2}Vr1*\frac{{Cr}\; 1a}{{{Cr}\; 1a} + {{Cr}\; 1b}}}}$Vdc 1 a = Vin/2 − Vdc 1 b

Where Vdc1 b is the dc voltage across Cr1 b, Vr1 b is the rms value ofthe ac part of the voltage across Cr1 b, Vr1 is the rms value of the acpart of Vr (the voltage across both Cr1 a and crib), Vdc1 a is the dcvoltage across Cr1 a, Vin is the value of input voltage, Cr1 a and Cr1 bare the corresponding capacitance of resonant capacitors Cr1 a and Cr1b. Due to the unidirectional current capability of the clamping diodeDc1, if the minimum voltage of the clamped capacitor is to be maintainedat zero, the following relationship should be maintained in steadystate:

Vdc1a/Rx1a≤Vdc1b/Rx1b

Rx1 b and Rx1 a are the parallel resistance across capacitor Cr1 b andCr1 a, correspondingly. If this relationship is not maintained, the dcvoltage across Cr1 b will be increased until this relationship is met.In such a case, the clamping diode Dc1 will not conduct any current insteady state operation, and theoretically can be removed. However, toprotect the circuit in transients, in a preferred embodiment, Dc1 canstill be kept as a safety measure. Cr1 a and Cr1 b do not have to beconnected directly. For example, Cr1 a (and its corresponding parallelresistor Rx1 a) can be moved across L1 without affecting the circuitoperation. The above analysis can be similarly made for the componentsin the receiver, or in any resonant circuit in general.

Variable capacitance can be implemented by combing the voltage-shiftingtechnique with voltage controlled variable capacitors, or morepreferably with switched capacitors as is shown in FIG. 12A. In thisfigure parallel resistors across Cx1 through Cx4 are not shown forsimplicity, and such parallel resistors can be implemented as theinternal parasitic resistors as explained earlier. The variablecapacitance configuration in FIG. 12A can be used to replace anyresonant capacitor whose capacitance needs to be controlled, such as Cr1and Cr2 in FIG. 7. The operation of the configuration shown in FIG. 12Ais similar to that of FIG. 8A. But with the help of clamping diode Dc,now Sx1 through Sx4 can be implemented as unidirectional devices such asa MOSFET, as is shown in FIG. 12C. This significantly reduces the costand power losses associated with the implementation of variablecapacitance.

It's better to turn-on or turn-off a switch when the resonant capacitorvoltage is at its lowest voltage to reduce the disturbance to thecircuit operation, and the stress to the switch. The switching time canbe determined by monitoring the resonant capacitor voltage or the coilcurrent, as the minimum voltage points coincide with the current zerocrossings in the positive direction.

This variable capacitance technique can be used in different topologies.FIG. 13 shows an example of a transmitter with Class E topology.Obviously, if necessary Cs can also be replaced by a variablecapacitance. However, due to the clamp effect of the body diode of S1,Cs sees only a unidirectional voltage, so voltage shifting is notnecessary.

Traditionally, in a resonant coupled WPT system, the resonant capacitorand the coil in the transmitter, as well as the resonant capacitor andthe coil in the receiver, are designed to resonate at the operatingfrequency. The inductance of the transmitter coil and that of thereceiver coil vary significantly due to coupling change when the coilsare placed at different locations, and both capacitance and inductanceof the resonant components have wide tolerances due to manufacturing andmaterial differences. To set the resonant frequency of the system, it isnecessary to adjust the capacitance or the inductance of the resonanttanks in the transmitter and receiver. It is usually complex to adjustthe inductance of an inductor in real-time operation. Therefore, it isdesirable to adjust the capacitance of the resonant capacitors toachieve acceptable performance. The above mentioned low-cost andefficient implementation of variable capacitance can serve such purposewell.

One particular important aspect in a resonant WPT system is how to setthe resonant frequency, as there are different resonant operation modes.In one embodiment, the resonant capacitor resonates with thetransmitting or receiving coil at the operating frequency (resonantcoupling technique), with the following characteristics:

-   -   The power transfer between the transmitter coil and the receiver        coil is maximized;    -   The voltage in a receiving coil heavily depends upon the mutual        inductance. Since the mutual inductance varies significantly        with the distance between the receiving coil and transmitting        coil and the orientation of each coil, the design is very        complex and usually such design has low efficiency;    -   Because high Q is necessary in the system and the system        operates at the resonance point, the operation of the system is        very sensitive, and components in the system usually suffer from        high stress;    -   At the resonant frequency, the resonant circuit has either very        low impedance (in a series resonant circuit) or very high        impedance (in a parallel resonant circuit). To interface such an        extreme-impedance circuit with the input source or the load,        impedance matching circuits are usually used. Such matching        circuits further increase the operation complexity and cost,        while reducing the system efficiency.

In an alternative embodiment, a WPT system can work in “leakageresonant” mode: the resonant frequencies resulted from the leakageinductance and resonant capacitance in both the transmitter and receiver(fr1=1/(2π√{square root over (Lx1Cr1)}) in the transmitting coil andfr2=1/(2π√{square root over (Lx2Cr2)}) in the receiver) are the same asthe operation frequency. In this mode, if S1 and S2 is operated with afixed duty cycle, the ideal ratio of the output voltage to the inputvoltage is determined by the physical parameters of the receiving coiland the transmitting coil (for example, number of turns of each coil),and is less dependent upon the coupling between the coils. This canresult in better performance and simpler design. With this operationprinciple, additional impedance match circuit can still be used tointerface a resonant tank to other part of the circuit, but it is nolonger necessary. Also, an optional filter circuit can be used tointerface the transmitter coil or the resonant tank in the transmitterto filter out higher frequency current, so that the energy transmittedby the coil has lower harmonic contents. The essence of this leakageresonant technique is to compensate the effect of leakage inductance ofa coil by a resonant capacitor, as the leakage inductance is especiallyprominent in WPT systems due to the poor coupling between thetransmitter and the receiver coils.

However, as will be made clearly later, it's not necessary to completelycompensate the effect of the leakage inductance for this technique to beuseful in a design. Sometimes it may be even better to change theresonant capacitance away from the leakage resonance point, i.e. haveonly partial compensation, to regulate the output voltage. The abilityto control the output voltage by changing the capacitance of theresonant capacitor in the receiver or the transmitter, or anycombination thereof, provides greater system control flexibility in WPTsystems.

FIG. 14 shows a block diagram of a WPT system consisting of resonantconverters with variable capacitance. In this system, the control ofoutput voltage is achieved through another control loop to adjust theinput voltage to the resonant converter in the transmitter, and thesignal transmission required can be accomplished through the modulationof the resonant power conversion (for example, through modulating Vin),or through additional communication channel such as a Bluetooth link. Acapacitance change control 1426 is used to determine the resonantcapacitance of a power converter 1425 in the transmitter, to optimizethe operation of resonant power converter, such as to fine tune theresonant frequency, to minimize resonant current to reduce the powerlosses, to minimize the voltages across the resonant capacitors, etc.Similarly, another capacitance change control 1456 can be used todetermine the resonant capacitance of a rectifier circuit 1425 forsimilar purposes. It's not necessary to use variable capacitancetechnique in both transmitter and receiver, although it may beadvantageous to do so as will be made clear later. Please note thatphysically the variable output power supply 1421 and related powercontrol 1422 can be put outside the transmitter 1420, and the outputcontrol 1452 can be also put outside the receiver 1450 physically. Ofcourse, it is also possible to include the load physically inside thereceiver.

The use of the variable capacitance technique proposed in this inventionwill improve the performance of the system in FIG. 14. But moreinterestingly, by changing the resonant capacitance in the transmitteror receiver, the output voltage can also be adjusted over certain range,without using a variable output power supply. Such a system is shown inFIG. 15A, which allows several control strategies to be devisedconsidering there are two control variables (the resonant capacitance inthe transmitter and the resonant capacitance in the receiver):

-   -   Change the resonant capacitance in the transmitter to control        the output voltage; In general, a lower resonant capacitance in        the transmitter gives higher output voltage, and a higher        resonant capacitance in the transmitter gives lower output        voltage;    -   Change the resonant capacitance in the receiver to control the        output voltage; In general, a lower resonant capacitance in the        receiver gives lower output voltage, and a higher resonant        capacitance in the transmitter gives higher output voltage;    -   Change resonant capacitance in the transmitter and the receiver        simultaneously to control the output voltage; the control in the        transmitter and the control in the receiver can have different        coordination schemes, for example:        -   a). The transmitter control provide a slower control, and            the receiver control provides a faster control, to regulate            the output voltage to a desired value;        -   b). The transmitter control provides feedforward control            against the input voltage change, input current change,            transmitter coil current change, or other parameter changes            in the transmitter circuit and any combination thereof,            while the receiver control regulates the output voltage to a            desired value using feedback control;        -   c). The transmitter control optimizes a circuit operation            parameter, such as efficiency, component stress etc, and the            receiver control regulates the output voltage to a desired            value;

FIG. 15B shows that the output voltage decreases with the increase ofresonant capacitance in the transmitter for a particular resonantcapacitance in the receiver in a particular design. The output voltagecan be controlled over a wide range by changing the resonant capacitancein the transmitter. Zero-voltage switching of the transmitter switchescan be maintained over a wide range of transmitter capacitance.

FIG. 15C shows that the output voltage increases with the increase ofthe resonant capacitance in the receiver for a particular resonantcapacitance in the transmitter in a particular design. Zero voltageswitching is kept for the transmitter switches over a wide range. Zerocurrent turn-off is kept for the receiver diodes and switches (assynchronous rectifiers) under all conditions. Due to the soft switchingof all power switches (diodes) in both the transmitter and the receiver,a high efficiency can be achieved over a wide range.

The above figures also clearly demonstrate that it's not necessary tooperate a WPT system at the coil resonant frequency or leakage resonantfrequency to get good performance. Therefore, regulating the outputvoltage over a certain range with resonant capacitor capacitancechanging is a valid control strategy. Also, in systems with variablecapacitance, the output voltage for the same capacitance configurationdecreases when the load current increases. That is, the output exhibitssignificant output impedance in open-loop operation. On one aspect, thesystem can observe a load change by measuring or estimating suchinformation as load current, input current, transmitter coil current,receiver coil current, coil current phase relationship with switchtiming in transmitter and other operating parameters, and use suchinformation change the resonant capacitance in the transmitter or thereceiver to achieve feed-forward control of the output voltage. On theother aspect, such relationship can be intentionally used to optimizepower delivery to the load. For example, the voltage of a batteryincreases when its charge increases. To charge a battery, the chargingcurrent can be reduced as the battery voltage increases above certainvalue. This allows the charging system with the variable capacitancetechnique described above to be optimized for lower cost whilemaintaining a reasonable performance.

In addition to the voltage control function, the capacitance controlblocks 1506 in the transmitter and 1516 in the receiver shown in FIG.15A can also optimize the performance of the transmitter and thereceiver as discussed for FIG. 14, because there are two variablecapacitances: one in the transmitter and one in the receiver. If suchoptimization results in a situation that the output voltage losescontrol, then the feedback signal passed by the signal transmission linkcan be used to correct the situation. It is also possible to finish thevoltage control without any communication link between the transmitterand the receiver, as is shown in FIG. 16. The transmitter can have afeed-forward control mechanism against the input voltage change,resonant parameter change and even load change, as the load change inthe receiver is also reflected in the operation parameters of thetransmitter, including the coil current amplitude or phase againstswitching timing. The fine regulation can be done in the receiver, againby changing the resonant capacitance.

Also, it's possible to use the resonant capacitance control to protectthe transmitter or receiver in fault conditions. For example, it thereis fault condition detected in the transmitter or receiver, the resonantcapacitor can be intentionally set to a very low or very highcapacitance, so the resonant tank is worked at a frequency far from itsresonant frequency, and the power capability is thus limited. Similarly,a transmitter or a receiver can be put into idle mode by intentionallysetting the resonant capacitor to a very low or very high capacitance.Such protection scheme is independent of the control scheme or powerstage topology.

One prominent advantage of the systems in FIGS. 15 and 16 is that theinput voltage to the resonant power converter can be fixed, and novariable voltage power supply is necessary. This significantlysimplifies the system design and reduces system costs. However, thecapacitance control scheme can also work well with a system withvariable voltage power supply and additional control loops, as is shownin FIG. 17. One design consideration for such a system is that the speedof different control loops should be coordinated accordingly.

One interesting strategy is to just roughly regulate the output voltageinto a voltage band without precise control. For example, the outputvoltage of a WPT system can be fed into a USB power input for a portableelectronic device, and the internal voltage regulation circuit in theportable device can regulate the USB port voltage into a more stablevoltage and/or charge a battery. In such cases, the leakage resonantfrequency of the transmitter and the receiver can be tuned to theswitching frequency so the output voltage will stay about the sameregardless of load condition. With a proper design of the transmittingcoil and the receiver coil, the right voltage can be achieved for suchapplications without a complex control scheme such as feedbackregulation. This can be considered as a WPT bus converter.

So far all the discussion is based on a single receiver coupled to asingle transmitter. The technologies are also applicable to systems withmultiple receivers, multiple transmitters, or any combination thereof.FIG. 18 shows an example with two receivers 1820 and 1830 coupled to atransmitter 1810. All control concepts discussed above can be used.

In many power converters, it's important to get accurate currentinformation. It is desirable to get current information from the powerswitches directly. Traditionally such sensing scheme is applied toMOSFET switches. However, because the switch current is usually achopped waveform of a continuous current, its frequency bandwidth ismuch higher than the original continuous waveform. Therefore, it's moreadvantageous to sense the combined currents from both top switch S1 andbottom switch S2 in a totem pole configuration, as shown in FIG. 19. Inthis figure, MOSFETs S1 and S2 form a totem pole configuration 1910,which can be used in many different converter topologies such as buck,boost, buck-boost, half-bridge, full-bridge, etc. In practice, powerMOSFETs are designed with many basic switch cells in parallel. Sense FETS1 s uses the same design of switch cells as in S1, but has much lessnumber of switch cells. By keeping the gate voltage of S1 s the same asthat of S1, the current in S1 s is a scaled-down representation of thecurrent in S1, with the scaling factor Ks being the ratio of numbers ofbasic switching cells in S1 and S1 s. Similarly, S2 s can sense thecurrent in S2 with a scaling factor determined by the numbers of basicswitch cells in S2 and S2 s. S1 s and S2 s are configured into a similartotem pole 1920 as 1910, with the mid-point of the totem pole fed intoan operational amplifier IC1 in signal processing block 1930. If thescaling factors of S1 s and S2 s are the same, then the current flowsthrough R2 is a good representation of Ib with the same scaling factor,Ib being the current in the totem pole. If the current gain oftransistor Tr1 is β, which can be made much higher than 1, then thecurrent flows out of the emitter of Tr1 will be

${Isense} = {\frac{{Ib}\left( {1 + \beta} \right)}{Ks\beta} \cong {\frac{Ib}{Ks}.}}$

Isense has the same spectrum as in Ib, thus the bandwidth and slew raterequirements of IC1 is much reduced compared to in the situation ofsensing the current of one switch.

If Ib is of negative value, Tr1 will cause an error as it cannot processnegative current as configured in FIG. 19 unless IC1 has also negativebias. The scheme in FIG. 20 can be used to overcome such limitation. Inthis scheme, a current source I1 is added to Tr1, so its current can bemaintained positive all the time. To cancel the effect of this addedbias, another current source 12, which can track I1 very well with stateof art IC design technique, is added, and both signals are processed bya differential amplifier IC2. Therefore, the output voltage of IC2 is agood representation of Ib. As well known in the industry, techniquessuch as switching edge blocking can be used to reduce switching noise inthe sensed signal. Temperature compensation and calibration can be usedto increase the accuracy.

As well known in the industry, use of synchronous rectifiers in thereceiver can improve the efficiency, and also reduce the voltage changewhen load changes, especially at light load. In a WPT system, thesynchronous rectifiers should be controlled with the current informationin the receiver coil, as it's difficult to get real-time switch statusinformation from the transmitter. To imitate ideal diodes, a MOSFET isparalleled with diode a (usually its body diode) and is gated ON whenthe diode (usually the body diode) conducts due to positive currentthrough it. When the current through it becomes negative or less than athreshold, the paralleled MOSFET is gated off. To reduce the effect ofnoise in the high frequency environment, an interlock mechanism shouldbe applied to complementary switches.

To get high efficiency at high frequency, it's necessary to take allmeans to reduce the switching loss of power switches. In a WPT system,usually the power switches in the transmitter can operate withzero-voltage turn-on, so more attention should be paid to the turn-offloss. A snubber capacitor can be put across a power switch to reduce thevoltage rising speed at turn-off, and thus reduce the turn-off loss. Dueto the use of zero-voltage turn-on, such snubber capacitor will notincrease turn-on loss. To reduce the effect of parasitic inductanceassociated with packaging and connecting traces, it's better tointegrate the snubber capacitor and the switch in the same package,preferably in the same die. Also, a gate drive with a negative turn-offvoltage can be employed. FIG. 21 shows a circuit diagram to create anegative turn-off voltage. In this drawing S is the power MOSFET to bedriven. The switch network 2111 consisting of Son and Soff takes turn-onsignal ON and turn-off signal OFF and creates a pulsed predrive signalVdrv. Although shown as MOSFET, Son and Soff can also use otherswitching devices, such as bipolar transistors (BJT). A waveform shapingcircuit 2112 can adjust the dc voltage and ac voltage differently. Cg isthe capacitance between the gate and source of MOSFET S and includes theinternal parasitic capacitance. Resistors R1 and Rg form a voltagedivider to set the dc voltages across C1 and Cg. With proper values ofR1, Rg, C1, and Cg, a negative voltage can be applied across thegate-source capacitor Cg at turn-off. FIG. 22 shows the simulatedwaveforms. As can be seen, even though the predrive signal Vdrv has nonegative voltage (the reference is the source of MODFET S), the gatevoltage of S is negative when it is off. The application of a negativevoltage during turn-off also allows a device with lower thresholdvoltage to be used as the power MOSFET, giving more freedom to optimizethe performance.

For high frequency operation, the driver circuit shown in FIG. 21 can beintegrated in the same package (preferably in the same die) as the powerMOSFET S.

Significant power loss occurs during the voltage rise time when a switchis turned off. A positive feedback technique can be used to shorten thevoltage rise time, and thus reduce the associated power loss. FIG. 23shows the principle. A positive feedback network 2310 sense the voltagerise of power MOSFET S, and the sensed signal is used to turn on theturn-off switch Soff, creating more discharging current for the gate. Asa result, the turn-off process of the MOSFET S is strengthened. In oneembodiment, the feedback circuit 2310 is implemented as capacitor Cd anda resistor Rd. This positive feedback technique also helps to avoidfalse turn-on during the turn-off process caused by the miller effect ofthe MOSFET. It can be used in combination with other drive techniques tocreate desired drive waveforms.

With the above disclosed techniques, efficient power conversion can beachieved at high frequencies. In actual design, integration can improveperformance and reduce costs. In high frequency design, it is importantto integrate as many function blocks together in the same package or inthe same die as possible, in order to avoid the adverse effects ofpackage parasitics. Both the receiver and the transmitter can benefitfrom the integration. Generally, the drive circuit and the optionalsnubber circuit should be integrated with the power MOSFETs in the samedie. For a WPT system, the power MOSFETs, the clamp diode and thecapacitor switches can all be integrated into one device. FIG. 24 showssuch an example implementation, in which switch network S1, auxiliaryswitch network 24200, and diode Dc are integrated. In one embodiment,auxiliary switch network 2420 consists of multiple switches to implementa variable capacitance. With the advancement of semiconductortechnology, it is also feasible to integrate bigger capacitors into asemiconductor dies now. So snubber circuit can be added to the mainpower switches S1 and S2. FIG. 25 shows an example, in which snubbercircuit 2530 for S1 and S2, and dc link capacitor Cdc are integratedwith switch network 2510, which helps to contain the switching currentsof S1 and S2 locally. Although not drawn out explicitly, all or some ofthe capacitors in the variable capacitance of FIGS. 12A, 12B, and 12C,together with the associated parallel resistors, can be integrated too.To ensure performance at high frequencies, the gate drive circuit forthe switches should be included too. Moreover, a more completeimplementation can be obtained by integrating gate driver, currentsensing, variable capacitance control, capacitance optimizer, bias powersupply, system control and protection functions, as is shown in FIG. 26.The bias power supply can provide control power to the device, and alsoto outside components such as a blue-tooth chip or systemmicrocontroller. Output over-voltage protection may be needed as theability to control output voltage to a lower value is limited withvariable capacitance when the load is very low. Measures such as a Zenerdiode, a linear regulator etc can be utilized to limit the outputvoltage to an acceptable value if needed. If a short-circuit conditionat the receiver output is detected, the resonant capacitance in thereceiver can be set to a high value, and the resonant capacitance in thereceiver can be set to the low value, so that the currents in thereceiver and in the transmitter will be limited. Alternatively, theswitches in the transmitter can be disabled to stop power transfer insuch conditions. The system can go back to normal operation after theshort-circuit condition is cleared. Similarly, during a power-up theresonant capacitance in the transmitter should be set to a high value,and the resonant capacitance in the receiver should be set to a lowvalue. If no receiver is detected in the transmitting range of thetransmitter, the resonant capacitance in the transmitter can be set to ahigh value to limit the surge current when a receiver is brought intothe transmitting range. These system control function can improve theperformance of the system. To achieve best power conversion performance,the active components and some of the passive components related topower conversion can be integrated into a single die. System functions,such as beacon and signaling circuit, can also be integrated with thepower conversion circuit in the transmitter and receiver. In someapplications, all components for a WPT transmitter can be integrated ina single die or on the same substrate to make a complete module, and allcomponents for a WPT receiver can be integrated in a single die or onthe same substrate to make another complete module. Due to the largesize, the transmitting coil and receiving coil can be put outside themodules. Of course, it's not necessary to implement all these functionsfor a device to be useful, and some functions can be optional for aparticular design.

Although embodiments of the present invention and its advantages havebeen described in detail, it should be understood that various changes,substitutions and alterations can be made herein without departing fromthe spirit and scope of the invention as defined by the appended claims.

Moreover, the scope of the present application is not intended to belimited to the particular embodiments of the process, machine,manufacture, composition of matter, means, methods and steps describedin the specification. As one of ordinary skill in the art will readilyappreciate from the disclosure of the present invention, processes,machines, manufacture, compositions of matter, means, methods, or steps,presently existing or later to be developed, that perform substantiallythe same function or achieve substantially the same result as thecorresponding embodiments described herein may be utilized according tothe present invention. Accordingly, the appended claims are intended toinclude within their scope such processes, machines, manufacture,compositions of matter, means, methods, or steps.

What is claimed is:
 1. An apparatus comprising: a first switch networkcomprising a plurality of power devices coupled to a first port having afirst voltage; and a first resonant tank coupled to the first switchnetwork, wherein the first resonant tank includes a first coil and afirst variable capacitance network comprising a first capacitor inseries with a first variable capacitor network, and wherein the firstvariable capacitor network comprises a voltage shifting diode inparallel with a plurality of capacitor-switch branches, and wherein eachcapacitor-switch branch comprises a capacitor connected in series with acontrol switch.
 2. The apparatus of claim 1, further comprising: acontroller configured to adjust a capacitance of the first variablecapacitance network by switching on or off the control switches.
 3. Theapparatus of claim 2, wherein: the controller is configured to adjustthe capacitance of the first variable capacitance network in response toa change of the first voltage, a change of a current in the first coil,or a change of a voltage in the first variable capacitance network. 4.The apparatus of claim 1, wherein: the first coil is configured to bemagnetically coupled to a second coil, and wherein the second coil and asecond variable capacitance network form a second resonant tank coupledto a second port with a second voltage through a second switch network.5. The apparatus of claim 4, wherein: the first voltage and the secondvoltage are an input voltage and an output voltage of a wireless powertransfer system respectively, and wherein a capacitance of the firstvariable capacitance network and a capacitance of the second variablecapacitance network are controlled to regulate the output voltage andimprove an operation parameter of the wireless power transfer system. 6.A method comprising: magnetically coupling a first coil of a wirelesspower transfer system to a second coil of the wireless power transfersystem, wherein: the first coil is coupled to a first resonant tankcomprising a first variable capacitance network; and the second coil iscoupled a second resonant tank comprising a second variable capacitancenetwork; and adjusting a capacitance of at least one of the firstvariable capacitance network and the second variable capacitance networkto regulate an output voltage of the wireless power transfer system. 7.The method of claim 6, further comprising: adjusting a capacitance ofthe first variable capacitance network or a capacitance of the secondvariable capacitance network to improve an operation parameter of thewireless power transfer system.
 8. The method of claim 6, furthercomprising: adjusting a capacitance of the first variable capacitancenetwork through feedforward control in response to a change of an inputvoltage or a change of an output current of the wireless power transfersystem.
 9. The method of claim 8, wherein: the change of the outputcurrent is detected through a change of a coil current, a change of acapacitor voltage of a capacitor in one of one of the first variablecapacitance network and the second variable capacitance network, or achange of a phase of a current flowing through one of the first resonanttank and the second resonant tank.
 10. The method of claim 6, furthercomprising: adjusting a capacitance of one of the first variablecapacitance network and the second variable capacitance network to limita current or a stress of the wireless power transfer system in aprotection process.
 11. The method of claim 6, further comprising:adjusting a capacitance of the first variable capacitance network and acapacitance of the second variable capacitance network simultaneously toregulate the output voltage of the wireless power transfer system. 12.The method of claim 11, further comprising: applying a slow controlmechanism to the wireless power transfer system through adjusting acapacitance of the first variable capacitance network; and applying afast control mechanism to the wireless power transfer system throughadjusting a capacitance of the second variable capacitance network. 13.The method of claim 11, wherein: the first coil is a transmitter coil,the second coil is a receiver coil, and wherein a feed-forward controlmechanism is applied to the wireless power transfer system throughmodulating a capacitance of the first variable capacitance network; anda feedback control mechanism is applied to the wireless power transfersystem through modulating a capacitance of the second variablecapacitance network.
 14. The method of claim 6, further comprising:coupling a first switch network comprising a plurality of power switchesto the first resonant tank; and operating the plurality of powerswitches to achieve zero-voltage switching.
 15. A system comprising: afirst coil of a power system magnetically coupled to a second coil of apower system, wherein: the first coil is coupled to a first resonanttank comprising a first variable capacitance network; and the secondcoil is coupled to a second resonant tank comprising a second resonantcapacitor; and a controller configured to regulate an output voltage ofthe power system through adjusting a capacitance of the first variablecapacitance network.
 16. The system of claim 15, wherein: the powersystem is a wireless power transfer system; the first coil is atransmitter coil in a transmitter of the wireless power transfer system;the second coil is a receiver coil in a receiver of the wireless powertransfer system; and a communication channel between the transmitter andthe receiver is configured to pass information to control powertransferring of the wireless power transfer system.
 17. The system ofclaim 15, wherein: the first variable capacitance network comprises avoltage shifting circuit and a plurality of adjustable capacitancecircuits, and wherein: the voltage shifting circuit comprises a firstcapacitor and a second capacitor connected in series, and a diodeconnected in parallel with the second capacitor; and the plurality ofadjustable capacitance circuits is connected in parallel with the secondcapacitor, and wherein each adjustable capacitance circuit comprises aswitch and a capacitor connected in series.
 18. The system of claim 15,wherein: the second resonant capacitor is a second variable capacitancenetwork, and wherein a feed-forward control mechanism is applied to thepower system through adjusting the capacitance of at least one of thefirst variable capacitance network and the second variable capacitancenetwork.
 19. The system of claim 15, wherein: a first switch networkcomprising a plurality of power switches is coupled to the firstresonant tank; and a power switch of the plurality of power switchesoperates under zero-voltage switching through adjusting the capacitanceof the first variable capacitance network.
 20. The system of claim 15,further comprising: a third coil magnetically coupled to the first coilor the second coil, wherein the third coil is coupled to a thirdresonant tank comprising a third variable capacitance network.